Department of Physics
State University of New York
Stony Brook, NY 11794-3800

ABSTRACT. Background, basic ideas, and recent development of a new family of ultrafast superconductor digital devices are reviewed. Possible applications of this new digital technology are discussed.

1. Introduction

Since the late 1960s it has become clear that several features of Josephson-junction integrated digital circuits may make them uniquely suitable for fast processing of digital information. These features include:
  1. Availability of superconducting microstrip transmission lines capable of transferring picosecond waveforms over virtually any interchip distance with a speed approaching that of light, and low attenuation and dispersion - see, e.g., theoretical analyzes by Kautz (1979) and Ekholm (1990) and experiments by Andratsky et al. (1990) and Polonsky et al. (1993b). It is also very important that these lines can be quite densely laid out (since spacing between the lines and their width are limited only by the available patterning technology), while having low crosstalk.
  2. Availability of Josephson junctions which can serve as ultrafast (picosecond) switches. Simultaneously, the junctions can be impedance-matched with the superconductor microstrip lines. This property allows one to ensure the ballistic transfer of the generated waveforms along the lines. Finally, even at these low impedances (R~10 Ohms) the Josephson junction power consumption P = V2/R is quite low because of low voltages employed (V<3mV, i.e. P<1 mW). As a result, chips with Josephson-junction integrated circuits may generate so little heat that they could be packed very closely (see e.g. Anacker (1980) and Anderson et al. (1983)), thus reducing interchip signal-transfer delays to fundamental limits, again determined by the speed of light alone.

As an additional factor, technologies of fabrication of the Josephson-junction integrated circuits may be considerably simpler than those of the present-day semiconductor (both silicon and gallium-arsenide) transistors with similar design rules, while providing quite acceptable RMS. deviations of the main parameters.

These factors were responsible for much work in this field during the last two decades, exemplified primarily by the American IBM project (1969-83) - see Anacker (1980) - and the Japanese MITI project (1981-91) - see e.g. Kroger (1986). Unfortunately, these big projects did not result in a practical superconductor digital technology. I believe that the main reason for that failure was an unfortunate choice of circuitry, which concentrated on various types of "latching" (or "voltage-state") logic.

Figure 1

Figure 1. (a) The buffer stage in Josephson-junction digital circuits and schemes of its operation in (b) latching logic and (c, d) RSFQ logic. After Likharev and Semenov (1991).

Figure 1a shows a simplified version of the simplest logic gate, the buffer stage, used in latching logic. It employs a tunnel Josephson junction which is naturally underdamped and thus exhibits a hysteretic dc I-V curve (Fig. 1b). The junction is biased by a dc current Ib slightly lower than the critical current Ic. Initially the junction is in its superconducting state (V = 0, point 0 in Fig. 1b). An arriving signal current Iin drives the total junction current beyond Ic, and triggers its switching to its resistive state 1 with V = 2D(T)/e, so that a considerable part of the current (Iout) is steered into the load R (typically, through a microstrip line); the latter current serves as an output signal. In most latching logics - see, e.g., Hayakawa (1983), Gheewala (1982), and Hasuo (1993) - the current Iin is also used to suppress Ic simultaneously, but this fact changes nothing essential in our discussion. This 0 -> 1 switching process can be very fast, down to almost one picosecond - see Kotani et al. (1988).

However, the reset (the 1 -> 0 switching) cannot be achieved by merely turning the signal Iin off: the circuit remains in its state 1. The only practical way to reset the gate to its state 0 is to switch off the bias current Ib. In the latching logics, this periodic reset of all gates is achieved by using an rf rather than a dc current supply for all the gates; this waveform performs also a global synchronization of the whole device. Unfortunately, this operation mode has severe drawbacks:

  1. Practical generation of the necessary rf supply presents something of a problem - see, e.g., Arnett and Herrell (1980). In particular, the rf currents necessary to drive a VLSI circuit would be very high and cause considerable crosstalk.
  2. An undesirable effect (so-called punchthrough - see, e.g., Likharev (1976)) of the junction switching to the opposite resistive branch of its symmetrical I-V curve (Fig. 1b) during decrease of the bias current, limits the rf supply (and hence clock) frequency to a few GHz.
  3. The rf power required per gate is well beyond that which can be produced by a similar gate. Thus, the clock signals must be generated externally, so that the latching circuits are restricted to external (global) timing.

As a result of these drawbacks, no prospects have been found to increase clock frequencies of latching-logic circuits beyond a few GHz. This speed is higher than, but comparable to, those of the fastest semiconductor digital circuits which do not require helium refrigeration. This marginal advantage is probably insufficient to warrant commercial introduction of this digital technology. This is why recently much attention was turned to alternative "flux-state" (or "Single-Flux-Quantum", or "SFQ") logic which uses coding of the binary information not by the dc voltage, but by single quanta of magnetic flux (F0 = h/2e = 2.07x10-15Wb). SFQ devices can be divided into two big groups, defined by the method used to pass the information between logic circuits.

In static SFQ circuits, suggested and analyzed by Anderson et al. (1971), Fulton and Dunkleberger (1973), Likharev (1976), Hurrell and Silver (1978), Likharev (1982), Loe and Lee (1985), Likharev et al. (1985 b), Loe et al. (1988) and others, the information is passed in the form of dc flux (or supercurrent). These devices are of a high fundamental interest because of their capability to implement the reversible processing of digital information - for details, see Likharev (1982). However, in static circuits the intergate distance is severely limited (practically, to the nearest neighbors) by the inductance of the interconnects. A second disadvantage of this approach is the necessity of an rf supply/clock, with resulting limitations on speed, similar to those for a voltage-state logic. Finally, a detailed analysis shows that parameter tolerances in static SFQ circuits are forbiddingly low.

In dynamic SFQ circuits, first discussed by Nakajima et al. (1976), Nakajima and Onodera (1978), Hurrell et al. (1980), Hamilton and Lloyd (1982), Silver et al. (1985), Nakajima et al. (1983), Oya et al. (1985), and some other authors, information between logic devices is passed ballistically, along either passive microstrip lines or active Josephson transmission lines, in the form of very short (picosecond) "quantized" voltage pulses V(t) with the fixed area

Int V(t)dt = F0 = 2.07 mV-ps.

The essence of this idea is that these "SFQ" pulses can be quite naturally generated, reproduced, amplified, memorized, and processed by elementary circuits comprised of overdamped Josephson junctions. This unique ability, fully realized in some analog devices based on the Josephson effect, was virtually neglected in latching logic; moreover, in the latching logic circuits the SFQ pulse generation is an inherent reason for the punchthrough effect which limits operation speed.

Over two decades, from the mid-1960s to the mid-1980s, several suggestions on how to use SFQ pulses for processing of digital information and analog-to-digital (A/D) conversion were put forward by the authors cited above, as well as by Clark and Baldwin (1967), Anacker and Zappe (1972), Likharev (1974), Lum et al. (1977), Zappe (1974 and 1975). It was only in 1985-86, however, that a complete family of dynamic SFQ circuits, with the nickname RSFQ (standing for Rapid Single-Flux-Quantum devices) was suggested by a Moscow State University group - see Likharev et al. (1985 a) and Mukhanov et al. (1987). During the period 1985-91 this group, in collaboration with a group at the Institute of Radioengeneering and Electronics (IRE), then of the Soviet Academy of Sciences, designed, fabricated, and tested a few circuits containing several of the simplest basic components of the RSFQ family. These components were demonstrated to work at clock frequencies in excess of 100 GHz with quite decent parameter margins (see Kaplunenko et al. (1989 a and b), and Filippenko et al. (1991)), despite a rudimentary 5-um technology used for their fabrication. Simultaneously, numerical simulations had shown that transfer to a 1-um technology could increase the speed beyond the 300-GHz level - see Mukhanov et al. (1987, 1989, and 1991), Rylov (1991), Polonsky (1991), Kirichenko et al. (1991), and Kidiyarova-Shevchenko et al. (1991). Since 1991, the RSFQ idea has been adopted by several groups in the United States and other countries, and its development is moving rapidly.

An alternate family of dynamic SFQ devices was suggested, under the name "Phase Mode Josephson System", by Professor K. Nakajima and his collaborators at Tohoku University in Sendai, Japan - see Oya et al. (1985) and Nakajima et al. (1989 and 1991). This system used a single basic cell, the "ICF gate", and seemed much less flexible than the RSFQ family. I am not aware of any recent attempt to continue this approach.

This review paper is intended to give a brief review of RSFQ digital circuits. In contrast with the detailed review of the initial work by Likharev and Semenov (1991), I will emphasize more recent results (obtained by mid-1992).

2. Basic Components of RSFQ Circuits


The most elementary RSFQ circuit coincides with that shown in
Fig.1a, except that now the Josephson junction is overdamped, e.g., the tunnel junction shunted externally by a metallic resistor low enough to reduce its McCumber-Stewart parameter bc below 1. Figure 1c shows the dc I-V curve of the junction; in contrast to the underdamped case, the curve is single-valued. This implies that after a current pulse Iin(t) is over, the junction is automatically self-reset to its original superconducting state (V = 0). If we look at this picture more attentively, using equations of the Josephson dynamics - see, for example, Likharev (1986) - we will see something quite interesting. There exists a broad range of amplitude and length of the pulse Iin(t), within which it triggers a quantized jump of the Josephson phase f of the junction by Df = 2p - see Fig. 1d. This fact can be readily understood starting from the well-known analogy between the Josephson junction and the pendulum. In this analogy, biasing of the junction with the dc current Ib < Ic corresponds to applying a nearly critical torque to the pendulum, driving it to a position close to the critical angle fc = p/2. The short input-current pulse is equivalent to a kick which drives the pendulum beyond fc. If the pendulum is overdamped, the kick results in just one 2p-rotation, with its automatic reset to the sub-critical static state. According to the fundamental Josephson phase-to-voltage relation

df/dt= (2e/h)V(t),

such a "2p-jump" of f corresponds to generation of the SFQ voltage pulse across the junction. For typical present-day fabrication technologies, duration of the pulse is a few picoseconds, while the pulse amplitude is a few hundred microvolts.

If the dc bias current Ib is not too far from the critical value Ic, this SFQ pulse can be triggered by an incoming short pulse, with either the nominal or a somewhat different amplitude. It means that the circuit shown in Fig.1a can reproduce SFQ pulses, bringing their area Int V(t)dt to the nominal value F0, i.e., providing a moderate voltage gain if necessary. On the other hand, if the input pulse is too weak (e.g., presents digital "noise" due to parasitic crosstalk between the signal transfer lines) it is not reproduced by the circuit, so that the circuit also serves as a noise discriminator.

Figure 2a shows another key RSFQ circuit, the Josephson transmission line (JTL) comprising several Josephson junctions connected in parallel by superconducting strips of a relatively low inductance L<F0/Ic , and dc-current biased to their sub-critical state (Ib<Ic). After the 2p-jump of the Josephson phase is triggered in the left junction J1 by the input signal, the resulting SFQ pulse developed across J1 will trigger the 2p-jump in J2, and this process will continue until the pulse is reproduced at the right edge of the JTL. JTLs can also be used to amplify SFQ pulses (or, more exactly, to provide their current/power gain while conserving their voltage area). For that, critical currents of the junctions and the corresponding dc bias currents should grow in the direction of the pulse propagation, with the proportional decrease of the inductances.

If amplification of SFQ pulses is not needed, they can be passed along passive superconducting microstrip lines. In order to match these lines to other RSFQ components, one can use short segments of the JTLs together with matching capacitors (Fig. 2b). Recently, such circuits were used to demonstrate transfer of 5-ps SFQ pulses over distances up to 1 cm without noticeable attenuation - see Polonsky et al. (1993 b).

An evident generalization of the JTL (Fig. 2c) can be used to provide splitting of the SFQ pulse, i.e., reproduction of the input pulse A at each of its two outputs B and C, without noticeable decrease of the pulse voltage amplitude.

All these simplest circuits are reciprocal, and cannot be used for isolation, so one needs a buffer stage (Fig. 2d). In this circuit, critical current of the junction J2 is somewhat smaller than that of the junction J1. Now, if the initial pulse arrives from the circuit input A, it is applied to J1 alone, and triggers the 2p-jump of the Josephson phase in J1, leaving the phase across J2 virtually undisturbed. As a consequence, the SFQ pulse is reproduced and passed to the output terminal B. On the contrary, if a pulse arrives from the latter terminal, it triggers a current pulse in both J1 and J2. As Ic2 < Ic1 , the junction J2 reaches its critical state earlier, and performs the 2p jump. Hence, voltage across J1 remains close to zero, which means that the SFQ pulse does not reach the input terminal A. Figure 2e shows a useful recent synthesis of the JTL and the buffer stage, having larger parameter tolerances - see Polonsky et al. (1993a).

Figure 2

Figure 2. The simplest components of RSFQ circuits: (a) Josephson transmission line, (b) driver and receiver for transfer of SFQ pulses along a passive superconducting microstrip line, (c) SFQ pulse splitter, (d) buffer stage, and (e) one-directional JTL. After Mukhanov et al. (1987 and 1989).


In principle, the SFQ pulse can be generated by the circuit shown in Fig. 1a, by feeding the Josephson junction by a short non-quantized current pulse Iin arriving from, say, a semiconductor electronic device. A disadvantage of this circuit is that the pulse should be very short (e.g., a few picoseconds), and its duration should be within certain limits. A much less demanding way is to use a Josephson junction in parallel with the superconducting inductor L (i.e., the usual single-junction superconducting quantum interferometer), with the basic dimensionless parameter l= 2pIcL/F0 somewhere between 2 and 6 - see, e.g., Likharev (1974). In order to generate a single SFQ pulse, the interferometer may be fed by a usual dc current pulse, with only amplitude (but not length) within certain limits.

Figure 3 shows a more advanced version of such a DC/SFQ converter. If its input current I is increased beyond a certain threshold value Iup , the critical state of the junction J3 is achieved, and the SFQ pulse is generated across it. Simultaneously, the three-junction interferometer (J1-J3, L1-L3) is switched into another flux state. In order to reset the interferometer into its initial state, the current should then be decreased below a value Idown at which the 2p-jumps are triggered sequentially in the junctions J1 and J2. The reset is accompanied by generation of SFQ pulses across these junctions, which do not penetrate into the output JTL. Numerical simulations and experiments by Polonsky et al. (1993 a) have shown that such a converter may have extremely wide parameter margins (up to 60%).

Figure 3

Figure 3. A possible structure for a DC/SFQ converter. After Polonsky et al. (1993 a).


Figure 4a shows another key component of virtually all RSFQ circuits. It is essentially the two-junction superconducting quantum interferometer ("dc SQUID"). If the inductance L of the interferometer is chosen so that its basic parameter l= 2pIcL/F0 is close to 10, and the dc bias current Ib is close to 0.8 Ic, the circuit has two symmetric stable stationary states which differ by the direction of the persistent current Ip = F0/2L circulating in the loop. In other words, one of these states corresponds to an additional single flux quantum trapped in the superconducting loop of the interferometer.

Let us suppose that the persistent current is circulating counterclockwise (binary 0), so that it sums with I in J3: I3 = Ib/2+Ip < Ic. If now the SFQ pulse arrives at the input S, it triggers the 2p-jump in J3, but not in J4 which carried a lower dc current I4 = Ib/2 - Ip. As a result of the jump, the cell is switched to its opposite state 1 with the clockwise circulation of the persistent current. It is evident that now the reset (the 1 -> 0 switching) can be triggered by the SFQ pulse arriving at the R terminal. Simultaneously, an SFQ pulse V(t) is developed across J2, which can serve as an output signal F. The auxiliary junctions J1 and J2 defend the SFQ pulse sources from the back reaction of the interferometer in the case of a "wrong" signal, for example, the S (set) pulse arriving during the state 1. In this case, junction J2 (rather than J3) switches; in other words, the incoming single flux quantum "falls out" of the circuit through J2 if the interferometer loop is unable to accept it. One can see that for SFQ pulses the circuit works exactly as a standard RS flip-flop (latch): SFQ pulses can be trapped by this circuit, so that the information about their arrival can be conveniently stored there in the form of static magnetic flux, and released when necessary in SFQ pulse form.

Figure 4b shows an SFQ analog of the T flip-flop (i.e., a single-bit stage of a binary counter). This circuit is fed by the input pulses from its single input. Each pulse is split and injected to both arms of the interferometer, so that it always triggers the circuit switching to the opposite state. Numerical simulations by Polonsky et al. (1993 a) have shown that for an optimized version of this circuit the dc-bias-current margins can be as wide as 37%; the experimental result by Kaplunenko et al. (1989 a) for a slightly different version was found to be 30%. Note that a conceptually similar device was proposed by Hurrell and Silver (1978) and implemented by Hamilton and Lloyd (1982) long before the development of the RSFQ logic family. However, due to the use of a resistor (instead of Josephson junctions J3 and J4) for injection of SFQ pulses, that device had extremely small parameter margins (about 5%) and could not be used in large-scale-integration circuits.

Figure 4

Figure 4. The simplest RSFQ latches: (a) RS flip-flop, (b) T flip-flop, and (c) T0 cell. After Mukhanov et al. (1987), Kaplunenko et al. (1989 a), and Likharev and Semenov (1991).

Figure 4c shows a modification of the T flip-flop (the "T0 cell"), which allows one to read out its contents (in the complementary code) by the additional SFQ pulse T. The quantizing loop now contains three junctions (J3, J4, and J6), but due to the corresponding choice of its parameters, it still has two stable states. If the loop is in its state 0, with counterclockwise circulation of the persistent current, then junction J6 is biased subcritically, pulse T triggers the 2p-jump of its phase, and the SFQ pulse is developed at the additional output S. (Note that this operation switches the loop from state 0 to state 1.) If, on the other hand, the loop is in its state 1 (with clockwise circulation of the persistent current), the bias in J6 is small, and pulse T leaves it in the superconducting state (inducing the 2p-jump in the junction J5 instead).


Figure 5a shows a possible structure for an "SFQ pulse monitor" which is a combination of the T flip-flop and SFQ/DC converter, producing a dc voltage at its output. Its heart is the interferometer (J1, J3, L) connected to an additional pair of the Josephson junctions (J5, J6) forming another dc SQUID. If the basic interferometer is in its state 0, the Josephson phase drop across junctions J5 and J6 is small, and this additional SQUID is in its superconducting state (zero dc voltage across J5). Switching of the basic interferometer to its state 1 leads to reduction of the critical current of the additional SQUID, and to its transfer to the resistive state (accompanied by continuous Josephson oscillations of junctions J5 and J6), i.e., to the appearance of a non-vanishing dc voltage V at the converter output. A similar converter can be nested on the RS flip-flop as well (Fig. 5b).

Such SFQ/DC converters have been repeatedly tested experimentally to operate with more than 30% parameter margins, in good agreement with simulation results - see, e.g., Kaplunenko et al. (1989 a and b), Filippenko et al. (1991), and Polonsky et al. (1993 a).

Figure 5

Fig. 5. SFQ/DC converters combined with (a) a T flip-flop and (b) an RS flip-flop. After Kaplunenko et al. (1989 a) and Likharev and Semenov (1991).


In order to proceed to digital operations with signals as unusual as picosecond SFQ pulses, one needs an explicit definition of what digital information is in RSFQ circuits. Such a definition was probably the most important conceptual step made by Likharev et al. (1985 a).

According to this concept, any RSFQ circuit may be considered as consisting of "elementary cells" (or "timed gates") operating as Fig. 6 shows. Each cell has two or more stable flux states. The cell is fed by SFQ pulses which can arrive from signal lines S1,..., Sn, and a clock (timing) line T. Each clock pulse marks a boundary between two adjacent clock periods by setting the cell into its initial state 1. During the new period, an SFQ pulse can arrive (or not arrive) at each of the cell inputs Si (Fig. 6b). Arrival of the SFQ pulse at a terminal Si during the current clock period defines the logic value 1 of the signal Si, while absence of the pulse during this period defines the logic value 0 of this signal. (Note that this convention does not require the exact coincidence of SFQ pulses in time; nor is a specified time sequence of the various input signals needed.) Each pulse can either change or not change the internal state of the cell, but it can not produce any immediate reaction at its output terminal(s) Sout. Only the clock pulse T is able to fire out the pulse(s) Sout corresponding to the internal state of the cell, predetermined by the input signal pulses which have arrived during this period. The same clock pulse terminates the clock period by resetting the cell into its initial state. Thus, an elementary cell of the RSFQ family is equivalent to a usual asynchronous logic circuit coupled with a latch (flip-flop) storing its output bit(s) until the end of the clock period.

Figure 6

Figure 6. RSFQ logic: (a) elementary cell and (b) signal sequence. After Likharev and Semenov (1991).

3. Implementation of Single-Bit Functions: A Few Examples

3.1. YES (D CELL)

In order to understand how the above convention works, let us come back to the RS flip-flop (Fig. 4a) and suppose that the input S is fed from a signal line, while the clock pulses are fed into input R. The clock cycle starts from the clock pulse, resetting the system back into its state 0. If no signal pulse S arrived during the current clock cycle, then the new clock pulse R would trigger the SFQ pulse across J1 rather than J4, and no output signal appears at the output F. However, if this concluding clock pulse was preceded by the signal pulse S (switching the interferometer to state 1), then the clock pulse would trigger the SFQ pulse across J4, which will serve as the output signal (simultaneously with the new resetting of the flip-flop into state 0).

We see that the RS flip-flop in this case works in accordance with the above definition of an RSFQ elementary cell, performing the function YES, i.e. reproducing the input signal S, with its time delay until arrival of the clock pulse. In other words, this circuit works as a 1-bit stage of a shift register. Such RSFQ registers have been tested successfully by Mukhanov (1993) at frequencies up to 60 GHz with parameter margins up to 30%.


Figure 7 shows a possible implementation of an RSFQ inverter. This circuit is also built around one interferometer (J2, L1, J3), but also includes an additional Josephson junction J1. In the initial state 0 the higher current is flowing through J2, while J3 carries virtually no current. This is why, in the absence of the input pulse W1, the next clock pulse would trigger the SFQ pulse in J1 rather than in J3, and this pulse would appear at the circuit output J1 (so that input 0 provides output 1). If the signal pulse W1 arrived, it would switch the interferometer into the state 1, with a higher current in J3. In this case, the next clock pulse would trigger the SFQ pulse across J3 rather than J1, the circuit would be reset, and no output pulse developed (i.e., input 1 yields output 0).

Figure 7

Figure 7. A possible implementation of RSFQ inverter. After Kidiyarova-Shevchenko et al. (1991).

This circuit was suggested by Kidiyarova-Shevchenko et al. (1991) and recently tested experimentally by Polonsky et al. (1993 a) and Goldobin et al. (1993). The tests have shown that the inverter can have a critical margin approaching 25%, and dc supply margin approaching 30%, in good agreement with results of numerical simulations.

This (very typical) RSFQ cell also was used recently in two similar experiments by Goldobin et al. (1993) and Polonsky et al. (1993 a) for estimating the probability of rare errors. For that case, the inverter output was connected to its clock input through a JTL line. When an SFQ pulse was injected into this closed loop (through some additional circuitry), it circulated permanently with a frequency about 15 GHz, thus providing dc voltage V1 ~8 mV across the junctions of the JTL. At the margins of the parameter window, rare errors were clearly visible as random jumps of the voltage up (to a dc voltage of about 2V1 corresponding to two SFQ pulses circulating in the loop) and down (to zero voltage corresponding to no SFQ pulses). Preliminary estimates show that these errors were due to external interference rather than fundamental thermal fluctuations in the Josephson-junction shunts, because of unshielded environment with only rudimentary filtering. Even under these conditions, however, not a single jump has been observed by Polonsky et al. (1993 a) in the middle of the parameter window during 6 hours of observation, thus indicating that the error probability was less than ~3x10-15/bit at T = 4.2 K.

This encouraging result should be compared with the much larger probability of 2x10-7/bit measured by Durand et al. (1992) at the same temperature in the resistively-coupled SFQ T flip-flop. This discrepancy can be readily explained by the fact that the resistively-coupled flip-flop had very narrow parameter margins, so that no operating point inside this small parameter window was really stable.

Figure 8

Figure 8. Equivalent circuit of an RSFQ OR cell. After Polonsky et al. (1993 a).

3.3. OR

Figure 8 shows a circuit suggested by Polonsky et al. (1993 a), which implements the 2-input OR function. It consists of two quantizing loops (J1, L1, J2) and (J5, L2, J6), each with two stable states. Input signals IN1 and IN2 can switch these interferometers from their initial state 0 (with larger currents flowing through junctions J1 and J5, respectively) to state 1 (with larger currents flowing through J2 and J6, respectively). If no inputs have come during the current clock period, the next clock pulse triggers SFQ pulses across Josephson junctions J4 and J8, which do not penetrate to the circuit output. If the input pulse IN1 alone has arrived during the clock period, and thus junction J2 carries high current, the clock pulse CLK triggers the 2p-jump of the phase in this junction. The generated SFQ pulse easily passes through the small inductance L3, triggers the SFQ pulse across J9, and passes to the circuit output through the superconducting junction J10 and small inductance L5. Simultaneously, the SFQ pulse is developed across J12. If the input pulse IN2 alone has arrived, a similar process will take part in the lower part of this symmetric circuit. Finally, if both IN1 and IN2 have arrived, both junctions J2 and J6 develop SFQ pulses simultaneously. These pulses trigger simultaneous SFQ pulses across J9 and J11, but only one RSFQ pulse appears at the circuit output, because these junctions are connected in parallel, as seen from the output (in which case, junctions J10 and J12 remain superconducting).

When fabricated and tested by Kwong et al. (1993), this circuit has shown very large parameter margins for each of the dc bias currents I1-I4; estimated margins for the dc supply as a whole were close to 30%, in reasonable agreement with results of numerical simulations by Polonsky et al. (1993 a).


We have seen that the RS flip-flop (Fig. 4a) itself presents a single-bit "D" cell of a shift register with destructive readout (DRO). Figure 9 shows the equivalent circuit of a cell which allows a non-destructive readout (NDRO) of its contents. Its quantizing loop (J1, J4, L1, J2) can be switched between its two stable states by write-in pulses W0 and W1. Setting pulse W1 triggers sequential 2p-jumps in junctions J2, J7, and J6, and triggers counterclockwise circulation of the persistent current in the loop. Reset pulse W0 triggers a 2p-jump in junction J4 alone, and thus restores clockwise circulation. Note that in neither case does an SFQ pulse penetrate to the NDRO output port OUT, though one can easily pick up the usual DRO output pulse from J2 or J7.

Figure 9

Figure 9. RSFQ NDRO memory cell. After Polonsky et al. (1993 a).

If the NDRO-initiating pulse arrives (from the input port denoted as CLK in Fig. 9) when the cell is in its state 0 with clockwise circulation of the persistent current, it triggers an SFQ pulse across J3, but leaves junction J1 (with small net current) in its superconducting state, thus producing no effect on the state of the cell. If, however, this pulse arrives when the cell is in the state 1 (counterclockwise circulation of the persistent current making the bias of J1 subcritical), it triggers 2p-jumps sequentially in J1 and J5, thus providing the output SFQ pulse to the NDRO output port. Note that for a moment the cell has changed its state from 1 to 0. However, the SFQ pulse across J5 passes through superconducting junction J6, small inductances L2 and L4, is reproduced across J7, and (passing through a small inductance L6) is fed back into the quantizing loop. It triggers a 2p-jump in J2 and thus returns the cell to its state 1 after a small (few-picosecond) delay.

This "delayed feedback" principle was successfully applied by Polonsky et al. (1993 a) to design several other circuits of the RSFQ family, including an RS flip-flop with two complementary outputs and demultiplexer. Numerical simulations and experimental testing of these circuits have shown that despite their relatively complex dynamics they can have quite decent parameter margins (about 30%).

Note that the NDRO memory cell can be considered as an asynchronous single-bit multiplier of signals W1 and CLK, provided that the former of these pulses arrives before the latter one.


A few other elementary cells including XOR (see Mukhanov et al. (1989 a)), AND (see Kirichenko (see Mukhanov et al. (1991)), and OR-AND (see Mukhanov et al. (1991)) cells, full and serial single-bit adders (see Mukhanov et al. (1987), Kidiyarova-Shevchenko et al. (1991), and Hamilton and Gilbert (1991)), and various multiplexers and demultiplexers (see Mukhanov et al. (1991) and Polonsky et al. (1993 a)), have been suggested, and some of them have been tested experimentally - see Polonsky et al. (1993 a) and Benz et al. (1993). General principles of their design are similar to those illustrated above. Now I shall discuss parameters of these cells.

  1. The number of Josephson junctions in a typical RSFQ cell turns out to be close to the number of p-n junctions in semiconductor transistor circuits implementing the same function.
  2. Power consumption of RSFQ cells is determined not by energy dissipation inside the Josephson junctions (which is typically as low as ~10-18 joule/bit), but by dissipation in dc supply resistors, and is of the order of 1 microwatt per gate, i.e., considerably less than that for latching logic.
  3. Depending on their definition, parameter margins for optimized RSFQ cells are typically between 20 and 30%. This implies that a reasonable fabrication yield of the VLSI RSFQ circuits could be achieved with the RMS scattering of the Josephson-junction critical current close to 5% (see Hamilton and Gilbert (1991) and Miller et al. (1993)), which seems a realistic target for present-day low-Tc fabrication technology - see, e.g., Tarutani et al. (1989) and Imamura and Hasuo (1992).
  4. But of course the most impressive feature of RSFQ cells is their extremely high operation speed, i.e., small logic delay. (For timed circuits such as RSFQ elementary cells, the delay is defined as the minimum value of the clock period for which the cell operates correctly.) Numerical simulations show that for a typical RSFQ cell, this delay is of the order of 3t0, where t0 is a natural unit of time:

    t0 = h/4eIcRs,

    where Rs is the external resistance shunting the junction with the critical current Ic. Due to condition bc = 1, t0 scales as (C/Ic)1/2, where C is the junction capacitance proportional to its area. Because Ic is determined by fluctuation stability of the RSFQ circuits (Ic ~100 uA for T = 4.2 K), t0 is determined solely by C, and hence by available patterning techniques. For standard niobium trilayer technology this unit expressed in picoseconds virtually coincides with the linear size of Josephson junctions expressed in micrometers. It means that for 3.5-um technology, t0 = 3 ps, and a typical elementary cell can operate at clock frequencies of the order of 100 GHz. If submicron technology is employed, external shunting may become unnecessary, and the operation frequency may approach 500 GHz (for unshunted junctions with C -> 0, t0 -> h/D(T), where D(T) is the energy gap of the superconductor used; for niobium, t0 -> 0.5 ps).

4. Timing

At this unparalleled speed, timing becomes an issue of primary importance. In particular, external (global) synchronization of VLSI circuits at frequencies beyond 100 GHz is relatively impractical. Fortunately, clock pulses for RSFQ cells are physically identical to the signal pulses, and hence can be generated inside RSFQ circuits. Thus, the external global synchronization can be complemented (and sometimes replaced) by very convenient local self-timing.

Figure 10

Figure 10. The simplest clock distribution systems: (a) "counterflow" and (b) "concurrent flow" of the clock (single lines) and data (double lines). After Mukhanov et al. (1989 a).

For relatively simple fragments of RSFQ circuits, self-timing schemes may be quite straightforward. Let us consider, for example, a shift-register-type structure (Fig. 10). (Here and below the data transfer will be denoted by double lines, and the clock by single lines). The first task for this structure is a single-step shift of all the data. It is easily achieved (Fig. 10a) by sending the clock pulse in the direction opposite to the desired signal propagation direction, with an appropriate time delay tc per gate:

tc, t > d,

where t is the clock period, and d is the maximum logic delay of the cell. Another simple timing method is shown in Fig. 10b. This scheme can work correctly only if the equation is satisfied, and the whole register is initially empty. In this case, a single clock pulse will trigger a motion of each single input bit along the whole register. Similar timing schemes can be used also for two-dimensional RSFQ structures - see Mukhanov et al. (1989 a).

For more complex circuits, however, another ("hand-shaking") approach may be preferable. In this approach (borrowed by Mukhanov et al. (1989 b) from high-speed semiconductor electronics), each fragment (cell or block) of the system is complemented by a special circuit which generates the clock pulses for its correspondents. Figure 11a shows such a circuit for a shift-register type structure, while Fig. 11b shows the "coincidence junction" circuit employed for the timing. The latter circuit generates the output SFQ pulse only when each of its two inputs have been fed by SFQ pulses.

Figure 11

Figure 11. RSFQ hand-shaking: (a) general scheme and (b) equivalent circuit of the coincidence junction C. After Mukhanov et al. (1989 b).

Let the register be filled by a string of data and all coincidence junctions have received their acknowledgment (ACK) pulses. The cell is waiting for the arrival of the SEND pulse signaling that the receiver cell is reset and hence ready to accept new data. This pulse triggers the coincidence junction which first produces the clock pulse T for its native cell and thus forces it to send its output signal to the receiver. Simultaneously, this pulse is duplicated as the ACK signal necessary to set up the coincidence junction of the receiver, and (after an appropriate delay) as the SEND signal for the next (sending) cell. As a consequence, the entire data string will eventually be shifted by one step to the right. With a different initial setting, the same clock distribution system can operate in the "load" mode similar to that of the circuit shown in Fig. 10b.

The hand-shaking approach allows one to design self-timing circuits (providing appropriate time delays) locally, being sure that they can be later united to an arbitrary LSI circuit without any problem for correct operation of the system as a whole. It also allows one to replace the clock generators by clock controllers which would automatically adjust their pace to that of the slowest fragment of the controlled circuit - for a detailed discussion, see Likharev and Semenov (1991).

In complex VLSI circuits it probably will be beneficial to combine high-speed local timing (i.e., asynchronous operation) of the circuit fragments with the slower global timing of the circuit as a whole, thus ensuring its compatibility with the usual synchronous operation of its semiconductor environment.

5. RSFQ Arithmetic: Two Examples


Figure 12 shows a possible structure of an RSFQ circuit performing serial multiplication of two multi-bit numbers A and B. The block uses three types of single-bit elementary cells: DRO cells D (e.g., simple RS flip-flops, Fig. 4a), NDRO cells N (Fig. 9), and full adders FA (operating in the carry-save mode, i.e., as the serial adders). Operation of the block is controlled by two sequences of the clock pulses, TA and TB.

Figure 12

Figure 12. A possible structure for a serial RSFQ multiplier. After Mukhanov et al. (1989 a).

Figure 13

Figure. 13. Parallel RSFQ multiplier: (a) general structure and (b) internal structure of the unit M (units D and H can be similar in structure, but perform only some of the functions of the unit M). After Mukhanov et al. (1989 a).

The operation is started by rapid loading of all n bits of number B into the cells - such loading can be provided, e.g., by the concurrent-flow timing scheme shown in Fig. 10b. Then the clock train TA starts, inducing gradual (one bit per clock cycle) loading of the number A into the shift register formed by the D cells. The latter train triggers also a simultaneous backward motion of the bits along the string of the full adders FA.

One may be readily convinced that at the end of each operation cycle (taking 2n clock periods) the circuit really does produce the correct 2n-bit product AxB at its output terminal P. (In fact, all the partial single-bit multiplications are performed by the NDRO cells, while the serial adders FA merely sum up all the partial products.)

Note that the circuit is occupied by the operands and the product during all 2n clock cycles, so that its throughput (number of operations per second) is relatively low - as an example, see Table 1 below.


The same operation can be carried out with much higher throughput using parallel-pipeline single-bit units. Figure 13 shows an example of such a device, a multiplier of two n-bit numbers A and B - in this example, n = 4. Note that the unit M is a copy of a single column of the serial multiplier (Fig. 12); the only difference is the method of connecting the units and their timing.

TABLE 1. Estimates of basic parameters of 32x32-bit fixed-point multipliers based on various digital circuit technologies. After Mukhanov et al. (1989 a)

Circuit type;
fabrication technology
Design rules
Integration scale
(103 Josephson
or p-n junctions)
(109 operations
per second)
Parallel-pipelined; Si-MOS 1.0200.00.2150
Parallel; JJ latching 2.570.00.52
Serial; JJ RSFQ
Parallel-pipelined; JJ RSFQ 2.540.030.02

If clock pulses are fed into the vertical columns of the structure sequentially, from right to left (for example, using the counterflow scheme shown in Fig. 10a, with proper precautions to avoid clock skew), the signal bit front moves from left to right, the data being processed simultaneously. If fed in parallel by all 2n bits of new operands (A, B) each clock period, this "pipeline" multiplier produces all 2n bits of the product P = AxB simultaneously, during one period. Thus, the throughput of this circuit may be extremely high (one number per each clock cycle), although a full processing of each specific pair of the operands takes 2n+1 clock periods. The price for this high throughput is hardware. An example of performance of these two RSFQ multipliers is given in Table 1.

One can see that serial RSFQ devices can combine operation at reasonably high speed with exceptional simplicity. At the same time, parallel RSFQ devices can provide unprecedented throughput. A flexible trade-off between these two performance factors is also possible. For example, one can increase the speed of the serial multiplier by using parallel m-bit multipliers (m < n) instead of single-bit multipliers. Such a device would require a factor of m more junctions, but would have an m-times larger throughput.

The most important feature of Table 1 is that RSFQ circuits can provide a considerable improvement (at least two orders of magnitude) in performance in comparison to present-day semiconductor digital technologies.

6. Possible Applications of RSFQ Technology


The simplest and hence the most immediate application of RSFQ technology is analog-to-digital conversion. The reasons for this are: the extremely high switching speed of the Josephson junctions (and hence very short aperture time ta of the converters) and the natural 2p-quantization of the Josephson phase f (i.e., the F0-quantization of the magnetic flux).

Two types of Josephson-junction A/D converters have been developed during the last decade: parallel and series ones - for a recent review, see Lee and Peterson (1989). The former converters are usually believed to be able to provide the largest signal bandwidths. However, they need simultaneous delivery of ultrafast sampling waveforms to each of their n samplers, and high-precision analog input signal dividers. Both factors do limit the effective aperture time, so that the best experimentally demonstrated performance of the converters - see the recent work by Bradley (1993) - is comparable to that of their semiconductor commercially-available counterparts. I believe that much better prospects exist for serial converters. A possible basis of these latter devices, the so-called ripple counter, was proposed more than a decade ago by Hurrell and Silver (1978). Its practical application was, however, delayed until a method could be found to carry out ultrafast read-out and preliminary processing of the digital output of the device. RSFQ circuits provide such a method.

Figure 14 shows a possible structure for an RSFQ A/D converter - see Rylov (1991) and Likharev and Semenov (1991). Its first part (an SFQ comparator) is essentially the two-junction interferometer (J1, J2, L) formed by lower junctions of two Josephson-junction pairs which are fed by SFQ pulse trains of very high frequency (say, 100 GHz). If the flux Fe = MI applied to the interferometer is increasing and exceeds a certain value Fup, the next SFQ clock pulse will trigger a 2p-jump of the phase in J2, and an SFQ pulse will enter (through a small inductance L2) the digital part of the device. If flux Fe is decreased below a (typically different) value Fdown, a similar pulse will be generated by the lower part of the circuit. The digital part of the device starts with the corrector which cancels the pulses in both channels if they are generated during one SFQ clock cycle. (This outcome leads to the correct result eventually, because the number of pulses coming from two channels will be subtracted in the following circuit.)

Figure 14

Figure 14. Serial (counting) RSFQ A/D converter: (a) comparator and corrector, and (b) decimal filter. After Rylov (1991) and Likharev and Semenov (1991).

Figure 14b shows the digital part of the converter, the decimation filter. The circuit is timed by a continuous train of SFQ pulses. Two consecutive counts from each channel of the comparator-corrector unit are averaged by the delay of each count by one clock period in a simple register cell D and by addition of its contents to the next count. These averaged signals arrive at inputs of a 4-bit reversible counter using T0 cells (Fig. 4c), so that the register contents (in the complementary code) can be destructively read out. After every two clock periods the contents of the counter are read out (destructively) into the similar, but six-bit, binary counter ("accumulator"), again with signal averaging by additional D cells. After every four clock periods, a similar operation is fulfilled with the contents of the latter accumulator, etc., until the reloading frequency is reduced to the Nyquist frequency 2Fmax of the signal. (Note that T flip-flops in the clock distribution line provide division of the clock pulse frequency by two at each stage.)

One can readily be convinced that each accumulator provides the correct differential digital code of the input signal (except that all odd stages yield this information in the complementary code). The farther right that the counter is, the larger is the number of output bits (by two bits per stage) and the lower is the output frequency (by a factor of two per stage). An analysis of the device operation - see Rylov (1991) - shows that the 25% least significant bits of the output present the quantization noise and should be neglected, while the remaining bits are correct. (It is interesting and important that this result is not affected by moderate thermal noise.) Thus the overall accuracy of the converter is given by the formula

e = 1/2n = (pFmaxta)-3/2,

where ta is the aperture time of the counter (Fig. 14a), of the order of 5t0. In other words, due to oversampling and averaging of the sequential counts, the A/D converter accuracy improves by 1.5 bits for every octave of reduction in the signal bandwidth Fmax. It means, for example, that for 2.5-um design rules, n~16 correct bits can be obtained for the signal bandwidth of 100 MHz. To the best of our knowledge, this unique performance cannot be challenged by semiconductor devices, either current or projected.

Note that the formula coincides with that for semiconductor single-stage sigma-delta modulators - see, e.g., Candy and Temes (1992). In Josephson-junction technology a direct implementation of the sigma-delta modulators is, however, possible - see Przybysz et al. (1993) and Xiao and Van Duzer (1993) - only with a substantial loss of signal-to-noise ratio, because of the absence of convenient analog integrators.


The effective digital filtering described above can be also used for development of digital SQUIDs which would combine the high sensitivity of analog versions with a much higher slew rate and a virtually unlimited dynamic range - see Rylov (1991) and Likharev and Semenov (1991)). Consider the simple example shown in Fig. 15. The usual analog dc SQUID (J1, J2, L1) senses the current flowing in the input coil L2 of its dc transformer, and produces proportional changes of its output dc voltage V. After low-pass filtering, this signal controls the SFQ-clock-driven comparator formed by junctions J3-J5. If the dc signal I is above a certain threshold, the clock pulse T triggers the 2p-jump in junction J4, which is then applied to the clocked inverter and the register cell D. As a result, an SFQ pulse appears at the direct digital output, and also is injected to the positive (bottom) arm of the feedback loop. After passing through the Josephson transmission line (J7, J9, J11), the single flux quantum is injected into the pick-up coil of the SQUID. The polarity of the injection ensures reduction of the input flux applied to the dc SQUID by bF0 , where b << 1 is the feedback factor. This injection of a single flux quanta would continue each clock period until the analog output V of the dc SQUID reduces the input current I of the comparator below its threshold. As a result, the junction J5 rather than J4 would be switched each clock period, leading to a supply of SFQ pulses to the reverse digital output and insertion of the one flux quantum per each clock period into the negative (top) arm of the feedback loop. Hence, the proper operation point (right at the threshold) is approached from either side.

Figure 15

Figure 15. A simple RSFQ digital SQUID. After Rylov (1991) and Likharev and Semenov (1991).

Performance of this on-chip negative feedback is generally similar to that implemented by Fujimaki et al. (1988) with an important exception that in the RSFQ version the clock frequency f can be much larger (beyond 100 GHz instead of 0.5 MHz). As a result, the slew rate s = bF0f can be as large as ~1010 F0/s even in this simplest version of the digital SQUID. (We have used a realistic value b = 10-1 which would make the additional quantization flux noise dF = bF0/f1/2 = 3x10-7F0/Hz1/2 of the device comparable with the intrinsic noise of the best practical dc SQUIDs.) More complex versions of the digital feedback using SFQ pulse frequency multiplication (see Semenov (1993)) can presumably increase the slew rate further without increase of the SQUID noise. The SFQ digital output of the device presents the time derivative of the signal, just as in the A/D converters considered above, so that its counting/filtering can be fulfilled by the device shown in Fig. 14b. Note that the digital SQUIDs like that shown in Fig. 15 have a virtually unlimited dynamic range, because the static input inductance of the device is infinite, and current in the pick-up coil is never large. (This fact also allows one to use large and/or remote pick-up coils.)

One also can use RSFQ circuits to implement low-frequency A/D converters with the same property, but with active (and virtually infinite) impedance. Such devices, suggested by Semenov and Voronova (1989), could be used as digital voltmeters with a precision in excess of 10 decimal digits.


The next most suitable application of this new technology is digital signal processing, because many algorithms in this field can be implemented with relatively small on-chip memory. A typical example of such processing is digital filtering - e.g., see Kung et al. (1985) - where a linear filter with virtually any transfer function can be constructed of standard second-order sections. Such sections can be readily implemented using RSFQ circuits, and can allow superfast digital signal processing speeds. According to the estimate by Likharev and Semenov (1991), a second-order section handling a continuous flow of 32-bit numbers would require some 12,000 Josephson junctions and could have a throughput up to 5x108 numbers per second, even if implemented in standard present-day 2.5-um technology.

Thus, RSFQ circuits could extend the well-known complex methods of digital signal processing from "acoustic" frequencies (tens of kHz) to "radio" frequencies (tens and hundreds of MHz) typical for bandwidths of radar, TV, and communication systems.


In contrast with digital signal processing, a universal von Neumann-type computer is probably the most difficult system for improving performance with RSFQ (or any other superfast) technology. The reason is that such a system relies on frequent data exchange between the processor and memory, with the exchange rate being limited by at least the speed of light (~100 ps per 1-cm distance), and by present-day chip packaging technologies (~1 ns for data transfer to/from the chip). Direct implementation of such a computer using the RSFQ logic and an existing superconductor memory - e.g., see Kurosawa et al. (1989), Nagasawa et al. (1989), and Tahara et al. (1991) - would give a system some 10 times faster than those achievable with other existing technologies - see estimates by Likharev and Semenov (1991). This advantage may not be large enough to justify transfer to superconductor technology, with its necessity for helium refrigeration.

Nevertheless, there are several factors which encourage me to estimate the situation as not completely gloomy:

  1. Superfast microprocessors with limited on-chip cache memory can be extremely useful for controlling real-time signal processors - see the previous section - and for broadening the critical path bottlenecks of semiconductor supercomputers.
  2. The problem of the standard von Neumann-type computer being too slow is not specific to RSFQ digital technology, but will be met by any technology approaching the 100-GHz-clock-frequency frontier; it is just that RSFQ circuits have arrived there first. Solutions to this problem should be sought in the development of new computer architectures which make full use of the unique operation speed of these novel logic/memory circuits, and also take into account the final speed of signal propagation on both the intrachip and interchip levels.
  3. Another prospective direction for research and development is packaging. If a way can be found to allow SFQ pulse transfer between chips, one could place an entire computer on a large-size micro-module ("superchip"), where all the memory could be accessed ballistically with the speed of light. In such systems one could make full use of the unparalleled throughput of parallel RSFQ circuitry (cf. Table 1), and unprecedented computing speed would become a reality.

8. Conclusion. RSFQ: Advantages and Problems

Josephson-junction RSFQ circuits can perform logic and arithmetic functions at extremely high (sub-terahertz) clock frequencies, just a few times lower than the maximum internal speed to = h/D(T) of the superconductors employed. These circuits seem to represent the fastest digital technology currently available. A list of other advantages of this technology includes:

  1. need for only a dc power supply;
  2. small power consumption virtually eliminating self-heating problems up to the VLSI level, at least for low-Tc superconductors. This fact also leaves some hope for the eventual transfer of this technology to temperatures ~30K, using high-Tc Josephson junctions - see discussions by Likharev and Semenov (1991) and Miller et al. (1993);
  3. natural self-timing which enables one to retain ultra-high operation speed in some important VLSI circuits, notably the digital signal processors.

These impressive advantages are not meant to imply that RSFQ circuits are free of problems, but we should distinguish fictional problems (advertised in some recent popular publications) from real ones. One of the claims was that one could not avoid the effects of the parasitic flux trapping in superconducting thin films, especially in the ground plane, of an integrated RSFQ circuit. However, the experience of teams that worked on the IBM and MITI projects show that this problem can be solved by fairly simple magnetic field shielding.

Another myth is that the low amplitude (a few hundred microvolts) and short duration (a few picoseconds) of SFQ pulses would make fast communications between RSFQ circuits and a semiconductor electronic environment impossible. In fact, the existing SFQ/DC converters can deliver several hundred millivolts of dc voltage at their outputs in just a few tens of picoseconds. There seems to be no conceptual problem in the development of special superconductor drivers which would raise the output voltage to ~10 millivolts in a few hundred picoseconds - e.g., see Suzuki et al. (1990) - and semiconductor amplifiers with similar speed capable of reliable readout of these signals.

Finally, a popular myth stated that RSFQ circuits were "unreliable." This claim was never exactly formulated, and is thus very hard to cope with, and I can only hope that recent measurements of the rare error rate (see Sec. 3.2 above) would put an end to this myth.

Dealing with real problems, the first one to consider is the necessity for liquid helium cooling of low-Tc Josephson-junction circuits. Despite recent progress in closed-cycle cryocooler technology - see, e.g., Kotani et al. (1991) - refrigeration may create serious problems for some potential users, and one needs to have a sizable circuit performance advantage in order to justify the related pain. I believe that properly designed RSFQ circuits and systems will be able to demonstrate such performance edge in just a few years.

Another major problem is the absence of large Josephson-junction memories; demonstrated RAM chips had shown a decent access time of the order of 500 ps, but only of a few-kbit size. I do not see anything inherently wrong with implementation of high-density Josephson-junction RAMs retaining nearly the same speed, but of course their development would require a large investment of effort and money. Right now these resources are not in sight; probably they could be obtained only when RSFQ technology proves its practical value on the level of unique small-scale (say, single-chip) devices.

Hopefully, this new digital technology will be able to survive its painful evolution and perform a real breakthrough in superfast processing of information.


Multiple discussions of the problems addressed in this paper with many colleagues are gratefully acknowledged. The author is grateful to Dr. H. Weinstock for careful editing of the manuscript and many useful suggestions. This work was partly supported by DoD within the framework of the University Research Initiative (AFOSR Grant # F49620-92-J-0508) and by DARPA through the Consortium for Superconducting Electronics (Prime Contract #MDA 972-90C-0021).


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